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Home | Alpha Telephone | Domain Names | Web Hosting | Get Traffic | xrEvidence | xrSoccer United States Patent
Start and warm up features for electronic fuel management systems An electronic control unit for regulating the air/fuel ratio of an internal combustion engine is disclosed. The electronic control unit has an open loop calibration for regulating air/fuel ratio that is corrected with a closed loop correction signal developed by an integral controller. The open loop calibration is a speed-density based schedule which is combined with corrections for special conditions to account for the operating parameters of the engine at any instant. Included in these special condition corrections are provision for start and warm-up features. The start feature includes an enrichment pulse generator whose pulses have a duration dependent upon the engine operating temperature and speed which are inhibited while the engine is above a predetermined RPM. The warm-up feature includes a time and temperature dependent enrichment which is linearly multiplied by an engine load factor. The warm-up feature further includes a time dependent holding level or delay from which the enrichment factor value decays during warm up and a feature to defeat this holding level above a certain engine operating temperature.
Attorney, Agent or Firm: What is claimed is: 1. An electronic control unit for the management of the air/fuel ratio of an internal combustion engine, said electronic control unit comprising: base calibration means for regulating the air/fuel ratio of the engine in response to sensed engine operating parameters indicative of the mass air flow and mass fuel flow inducted into the engine, said base calibration means regulating the air/fuel ratio by sensing one of said mass air flow and said mass fuel flow and calculating the other from a schedule of desired air/fuel ratios; and starting means for enriching the air/fuel ratio during engine cranking by a starter motor wherein said starting means is enabled by a start signal indicating that the starter motor is cranking the engine and wherein said starting means includes crank override means to inhibit said enrichment if the speed of the internal combustion engine exceeds a predetermined value in excess of the cranking speed of the starter motor. 2. An electronic control unit as defined in claim 1 wherein said crank override means includes: RPM signal means for generating an RPM signal indicative of the speed of the internal combustion engine; RPM threshold means for generating a threshold signal indicative of said predetermined inhibit value; and comparison means for comparing said RPM signal and said threshold signal and for generating a crank override signal to inhibit said starting enrichment if said RPM signal is greater than said threshold signal. 3. An electronic control unit as defined in claim 2 wherein: said RPM signal is a voltage whose amplitude is responsive to the speed of the engine. 4. An electronic control unit as defined in claim 3 wherein: said threshold signal is generated as the junction voltage of a pair of divider resistances connected between a voltage source and ground. 5. An electronic control unit as defined in claim 4 wherein: said comparison means comprises a differential amplifier having an inverting input and a non-inverting input and an output wherein if the voltage applied to said input terminals is unequal said output terminal will be at one of two levels depending upon which input is greater in magnitude. 6. An electronic control unit as defined in claim 5 wherein: said RPM signal is connected to the inverting input of amplifier and said threshold signal is connected to the non-inverting input. 7. An electronic control unit as defined in claim 6 wherein said starting means includes: a pulse generation means for generating an RST pulse signal comprising a plurality of fixed duration pulses whose frequency is dependent upon engine speed; said starting means further including crank pulse generation means for generating cranking pulses whose duration are indicative of the desired start enrichment, said crank pulse generation means generating the cranking pulses in response to receiving said RST signal. 8. An electronic control unit as defined in claim 7 wherein: said amplifier has an open collector output and the crank override signal is generated by grounding said speed dependent pulse signal through the output of said amplifier. 9. An electronic control unit as defined in claim 8 wherein said RPM signal means includes: a timing capacitor receiving said RST signal and charging to the peak voltage amplitude of the signal during its presence, said timing capacitor further having a discharge path causing a decay of the voltage during the absence of said RST signal such that the voltage to which the timing capacitor decays is an indication of the speed of the engine. BACKGROUND OF THE INVENTION 1. Field of the Invention The invention pertains generally to fuel management system having an open loop calibration which includes provision for special condition calibrations and is more particularly directed to special condition calibrations for starting and warm up enrichment. 2. Prior Art Electronic fuel schedulers or electronic control units for regulating the air/fuel ratio of an internal combustion engine are conventional in the art. These schedulers provide, from a calculation or electronic computation based upon the operating parameters of the engine, an air/fuel ratio that is considered substantially ideal for the instantaneous conditions sensed. The "best" air/fuel ratio at which the engine will operate under a given set of operational conditions is normally a tradeoff between the competing factors of driveability, emissions, and fuel economy. It is generally understood that richer air/fuel ratios are better for power and driveability, a substantially stoichiometric air/fuel ratio the most desirable for emissions, and lean air/fuel ratios the calibration that gives the best fuel economy. The schedule of desired air/fuel ratios for the electronic control unit can be derived from empirical tests of emissions, driveability, and economy tests and may include areas where the one criterion is more important than the others. For example, under urban or in city driving, conditions emissions are considered of importance because of the congestion of automobiles present in a small area and the amount of pollutants at these slow speeds while at highway or freeway speeds, economy would be the overriding factor of consideration. In addition, for passing or accelerations and to ease starting and warm up situations, power and driveability must be factored into scheduling. Any number of the various engine parameters may be sensed to calibrate the schedule of air/fuel ratios, but the most advantageous method is to measure mass air flow or mass fuel flow and calculate the other from the schedule. An air/fuel controller having a calibration based upon the speed of the engine and the density of the air as a measurement of mass air flow has been successfully provided by a U.S. Pat. No. 3,734,068 issued to J. N. Reddy on May 22, 1973. The disclosure of Reddy is hereby expressed incorporated by reference herein. Reddy discloses a base calibration pulse width that is a function of the RPM of the engine and manifold absolute pressure. The duration of the pulse width is used to regulate fuel flow to the engine based upon a schedule. This base calibration is an open loop control of the air/fuel ratio as the operating parameters of the engine are sensed by the controller and a control signal which is the fuel pulse duration is developed therefrom. If the air/fuel ratio schedule from which the control signal is calculated or the engine environment to which it is applied is different from the optimum design system, then the controller will not perform as required. The difference in engine environments are generally either because of manufacturing tolerances that change the response of the engine, or, as occurs with all mechanical devices, the ageing factor which is difficult to schedule. It is known in the art that to solve many of the problems faced by open loop fuel schedulers a closed loop integral controller may be effectively utilized. The controllers are termed "closed loop" because they sense the result of an actual air/fuel ratio change and develop a control signal based therein rather than calculate an air/fuel ratio change from a desired schedule as does the open loop controller. One of the most advantageous of these controller systems is based upon the bi-level output of an exhaust gas composition sensor which indicates whether a rich or lean air/fuel ratio charge has been combusted by the engine. The controller incrementally leans the air/fuel ratio during a rich indication of the sensor and incrementally enrichens the air/fuel ratio during a lean indication of the sensors, thereby causing the system to oscillate in a limit cycle about a desired air/fuel ratio. Illustrative of this type of controller is a U.S. Pat. No. 3,815,561 issued to Seity which is commonly assigned with the present application. The disclosure of Seity is hereby expressly incorporated by reference herein. In addition to the speed-density calculation, the basic open loop calibration for mass air flow can and is generally corrected for special conditions. One of the more important of the special condition calibrations to the open loop schedule should be the starting condition where it is known that a much richer air/fuel ratio than normal is used to insure the quick and facile initial operation of the engine. One fuel management system with a start calibration makes use of the cranking of the engine to develop an enrichment pulse for every engine revolution. The enrichment or length of the pulses is dependent upon engine temperature with substantial enrichment occurring at lower engine temperatures and decreasing to a minimum as the engine warms to operating temperature. A positive detection of the cranking condition, the starter solenoid engagement is generated to enable the pulses during its presence. While this system produces the quick and facile starts it was designed to accomplish, it does create an emission problem during starter overrun. This time period is the short increment after the engine has started but before the operator recognizes the start has occurred by releasing the starter solenoid. Even though relatively short in length, this period can cause significant HC, CO emission impact because the starting enrichment pulses are speed dependent and of a length designed to provide a substantially enrichened air/fuel ratio at cranking speeds of between 30-60 RPM of the engine. As the engine starts, its crankshaft rotational speed rapidly increases to approximately five to ten times cranking speed to where the engine can sustain operation at an idle range of 300-600 RPMs or fast idle (850 RPM) in the best case. During the worst case condition, and generally in the manner many operators start engines, the engine will be accelerated to a much higher RPM than idle before the operator recognizes a start and releases not only the starter solenoid but also the accelerator pedal which he has depressed fully in his effort to start the engine. It would be desirable to suppress the emissions caused by these high rotational speeds and the enrichened cranking pulse width without leaning out the fuel mixture before starting the engine. The fuel management system described above further has a warm up calibration which is a function of both time and engine temperature. The temperature dependent portion provides enrichment during this period based upon temperature only with a greater enrichment provided at colder temperature and less enrichment at warmer temperatures. This enrichment which is somewhat less than the starting enrichment, provides a smooth transition between the start calibration and the operational or base calibration. Since it is dependent upon engine temperature, its magnitude will only be that which is needed for satisfactory driveability. The other component of the warm up enrichment calibration is time and temperature dependent. Initially, this component has a hold level enrichment value whose magnitude is based upon engine temperature. The level is a maximum at colder temperatures and decreases to a minimum at warmer temperatures. From this holding level, the enrichment level decays with a time constant to zero after a delay. It has been found that this system provides unnecessary enrichment during the holding or delay time at higher engine temperatures that are close to operational conditions. It would therefore be desirable, if when sensing these conditions, to defeat the holding period and proceed directly to the decay portion of the time and temperature dependent component of the warm up calibration. A further advantageous modification to a warm up calibration is one for load. When the engine temperature is cold, more enrichment for high loads is necessary than at the same load for a fully warm engine. Generally, a colder engine and a high load will necessitate the most enrichment and a light load with a fully warm engine, the least. A U.S. Pat. No. 3,971,354 issued to Luchaco et al, discloses a load dependent warm up enrichment circuit that provides enrichment as a function of MAP. The enrichment as a function of load provided by the Luchaco circuit is a modification of a temperature dependent warm up calibration and does not provide correction for a time dependent component such as that included in the present system. Further, Luchaco provides the most load enrichment in the power operating region of the MAP curve as seen in FIG. 4 of that reference and does not produce significant warm up load enrichment at MAP values below these portions of the MAP curve. It has now been found that enrichment provided for warm up values at loads within the normal operating range (300-600 torr) of MAP values increases driveability and response of the engine. Moreover, a linearly increasing enrichment smooths the transisions between load values in this intermediate range where accelerations and decelerations are common. SUMMARY OF THE INVENTION The invention provides improved start and warm up calibrations for an electronic control unit. The improved calibrations increase the driveability and reduce emissions of an automobile whose air/fuel ratio is regulated by an electronic control unit including an open loop base calibration dependent upon scheduling a quantity of fuel for a detected amount of inducted air. The start and warm up calibrations modify the base calibration to provide varying amounts of enrichment in response to the start and warm up conditions. The start feature includes a pulse generation means for generating enrichment cranking pulses during the cranking sequence of engine before starting. The duration of the cranking pulses are preferably temperature dependent and of a frequency dependent upon the engine speed. The cranking pulses are enabled via a signal representative of the engagement of the starter solenoid. The start feature further includes crank override means to inhibit the cranking pulses if the engine speed exceeds a predetermined value. The attainment of an engine speed in excess of the cranking speed of the starter is indicative of an engine start where the engine may sustain its own operation. Emissions during starter overrun may be reduced in this manner by inhibiting the cranking pulses even in the absence of the release of the starter solenoid. In a preferred implementation of the start feature, the pulse generation means receives a signal that is a series of reset pulses at the rate of engine revolution and generates the cranking pulses in a timed relationship to these reset pulses. A comparitor means receives an RPM voltage signal which is representative of the RPM of the engine and compares it to a threshold voltage indicative of the predetermined RPM value necessary for the engine to sustain operation. When the RPM signal exceeds the threshold, the comparitor means will inhibit the reset pulses independently of the state of the starter solenoid. The warm up feature includes a first warm up current generator means for generating a first warm up current proportional to the desired warm up enrichment which is time and temperature dependent. The first warm up current has a temperature dependent holding level which starts to decay to a minimum value after a predetermined duration of time. Further included in the warm up feature is a second warm up current generator means for generating a second warm up current whose enrichment is inversely proportional to temperature. The warm up enrichment provided by the warm up feature is a combination of the first and second warm up currents modified by a holding defeat means which causes the first warm up current to decay from the holding level immediately when the engine temperature of the internal combustion engine is in excess of a predetermined temperature. The defeat of the holding level eliminates unnecessary enrichment when the engine is already warm enough for smooth operation without eliminating the decay portion of the first warm up current which produces the even transition between the warm up and operational conditions. Preferably, the holding defeat means comprises a defeat comparator which receives a signal indicative of the engine operating temperature and compares it to a threshold representative of the predetermined engine temperature at which the defeating action should occur. This temperature is preferably the normal operating temperature of the engine. The defeat comparitor in response to the engine temperature exceeding the threshold will cause a holding comparitor to switch directly from the holding level to the decay portion without any delay. The total warm up current is additionally modified for the load of the engine by a load enrichment means. The load enrichment means varies the total warm up current as a linear function of the instantaneous engine load a maximum at full loads. Preferably the load enrichment means comprise a multiplier means which receives a signal indicative of the intake manifold pressure which is directly related to engine loading. For manifold pressures in excess of a set value related to off closed throttle positions of the throttle valve (a predetermined engine load), the multiplier means modifies the total warm up current to increase enrichment linearly for increases in manifold pressure to a full load condition at wide open throttle. The load enrichment means provides significantly greater enrichment for engine loads in the normal operating range of manifold pressures (300-600 TORR) than previous systems which provide enrichment only in the power operating range of manifold pressure. Further, a linearly increasing function produces generally better driveability than has been heretofore accomplished by load dependent warm up enrichments. Therefore, it is a major object of the invention to provide improved start and warm up enrichment features for an electronic control unit. It is another object of the invention to provide a crank pulse override to substantially reduce emission levels during starter overrun. It is still another object of the invention to provide a holding level defeat feature for warm up enrichment that eliminates unnecessary enrichment when the engine temperature has increased to an operational level. Yet another object of the invention is to provide a time and temperature dependent total warm up current that is linearly enriched as a function of the load of an internal combustion engine. These and other objects, features and aspects of the invention will be more fully understood and better described if a reading of the following detailed description is undertaken in conjunction with the appended drawings wherein: BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1a is a pictorial, partially sectioned block diagram of an electronic fuel management system for an internal combustion engine constructed in accordance with the teaching of the invention; FIG. 1b is a functional block diagram of the electronic control unit for the fuel management system illustrated in FIG. 1; FIG. 2 is a detailed system block diagram of circuitry comprising the electronic control unit of the fuel management system illustrated in FIG. 1; FIG. 3a is a detailed schematic diagram of the speed sensing circuit illustrated in FIG. 2; FIGS. 3b through 3g are waveform diagrams of signals at various terminal points of the circuitry illustrated in FIG. 3a; FIG. 4a is a detailed schematic diagram of circuitry for the pulse width generation circuit illustrated in FIG. 2; FIGS. 4b through 4e are detailed waveform diagrams of signals at various terminal points of the circuitry illustrated in FIG. 4a; FIG. 5a is a detailed schematic diagram of circuitry for the pressure sensing circuit illustrated in FIG. 2; FIG. 5b is a waveform diagram of the MFS signal generated in FIG. 5a as a function of pressure; FIGS. 5c-d is a functional illustration of the PWS signal generated in FIG. 4a as a function of pressure; FIG. 6 is a detailed schematic diagram of the injector driver circuit illustrated in FIG. 2; FIG. 7a is a detailed schematic circuit diagram of the correction current combination circuit illustrated in FIG. 2; FIGS. 7b-7e are waveform diagrams of signals at various terminal points throughout the circuitry of FIG. 7a; FIG. 8a is a detailed schematic diagram of the cold cranking function circuitry illustrated in FIG. 2; FIGS. 8b-8e are waveform diagrams of signals at various terminal points of the circuitry of FIG. 8a; FIG. 9a is a detailed schematic diagram of the AB curve correction circuit illustrated in FIG. 2; FIG. 9b is an illustrative pictorial view of an enrichment schedule as a function of the A curve current signal generated in FIG. 9a; FIG. 9c is a illustrative pictorial view of an enrichment, schedule as a function of temperature for the B curve current signal as generated in FIG. 9a; FIG. 10 is a detailed schematic diagram of the triangular wave generator circuit illustrated in FIG. 2; FIG. 11a is a detailed schematic circuit diagram of the positive K correction circuit illustrated in FIG. 2; FIG. 11b is a three-dimensional surface diagram of an enrichment schedule as a function of A curve current, B curve current and linear positive K enrichment; FIG. 12 is a detailed schematic odiagram f the altitude compensation circuit illustrated in FIG. 2; FIG. 13a is a detailed schematic diagram of the acceleration enrichment circuit illustrated in FIG. 2; FIG. 13b through 13h are waveform diagrams of various signals taken at various terminal points in FIG. 13a; FIG. 14 is a detailed schematic diagram of the fuel pump and safety circuit illustrated in FIG. 2; FIG. 15 is a detailed schematic circuit diagram of the closed loop control circuit illustrated in FIG. 2; FIG. 16 is a detailed schematic diagram of the failure detect circuit illustrated in FIG. 2; FIG. 17a are illustrative designations for the connection pins and internal registers of the 8048 microprocessor illustrated in FIG. 15; FIGS. 17b through 17e are illustrative waveforms of the closed loop control circuit illustrated in FIG. 2; and FIG. 18 is a detailed schematic diagram of an alternate implementation of the closed loop control circuit illustrated in FIG. 2; FIGS. 19a through 19k is an illustrative flow chart of the program stored in the read only memory in the microprocessor illustrated in FIG. 15. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT FIG. 1 illustrates an engine 11 of the internal combustion type having an air/fuel ratio management system comprising an electronic control unit 13. The engine 11 has numerous sensors that develop electrical signals based upon the operating conditions of the engine and transmit them to the electronic control unit for generating air/fuel ratio control signals based upon the parameters sensed. Electrical control of the air/fuel ratio increases the precision of the regulation for the air/fuel ratio during the constantly changing load, speed, and temperature conditions of the engine. This precise control in combination with presently available catalytic converters that eliminate certain exhaust products is utilized to reduce noxious emissions of the engine while maintaining driveability and good fuel economy. The system control is based mainly upon an open loop calibration of an air/fuel ratio charge inducted from an intake manifold 15 into the combustion chamber 17 of the various cylinders through an intake valve 19 of the engine during an intake cycle. The air/fuel charge is compressed and ignited by a timed spark device 21 as is known in the art and then exhausted during an exhaust cycle through an exhaust valve 23 into an exhaust manifold 25. Only one cylinder operation has been shown in FIG. 1 for the purpose of clarity but the preferred embodiment will generally be described as applicable to an eight cylinder automobile as indicated by the designations 1-8 on the distributor cap in the figure. The present system can, however, be easily adapted to any multi-cylinder internal combustion engine including compression ignited engines. In the preferred embodiment the amount of air (mass air flow) for the inducted air/fuel charge is varied in accordance with the position setting of a throttle valve 27 controlled by the operator. The ECU 13 senses the amount of air inducted and applies the open loop calibration to that sensed amount to calculate the scheduled amount of fuel. Generally, the calibrated open loop air/fuel ratio is substantially stoichiometric or a ratio of approximately 14.8:1. This air/fuel ratio is considered to be the best all around air/fuel ratio for operating the engine at while factoring in the competing goals of driveability, reduction of noxious emissions and economy. The particular system shown then injects with a plurality of solenoid type injectors 29 (one for each cylinder) the amount of fuel that the electronic control unit 13 has calculated from the engine operating conditions into the incoming air stream of the intake manifold. From the above discussion it is evident that the ECU 13 could also measure the amount of fuel flow and then calculate the needed amount of air/flow from an air/fuel ratio schedule. Still further, the invention should not be limited to electronically controlled injectors as electronic carburetors or electronically controlled air valves could as easily be regulated. The air flow is mainly calculated by the ECU 13 from a manifold pressure sensor 500 connected to the intake manifold 15 via a conduit 31. This pressure sensor 500 outputs an analog voltage which is representative of the pressure that is found in the intake manifold 15 and which will vary according to load conditions and the position of the throttle valve 27. The other parameter needed for indicating the air flow into the engine is a signal developed by an engine speed sensor 33 which provides two pulsed outputs at the RPM of the engine. The engine speed sensor 33 preferably has two normally open reed switches located on opposite sides of the distributor shaft that are operative to sense the passing of a set of permanent magnets fixed on the shaft by closing. The speed sensor 33 will output two signals, one from each reed, RD1 and RD2, respectively that will indicate the passage of the magnets past the pickups and, therefore, the speed and relative crank position of the engine. In this configuration, signals RD1 and RD2 are generated every engine cycle and are 180.degree. out of phase. From this basic speed density information representing the inducted mass air flow, the electronic control unit 13 will apply the open loop calibration to generate fuel pulse signals IG1 and IG2 to the injectors 29 and thereby generate the particular scheduled air/fuel ratio, such as stoichiometric. For the preferred eight cylinder example signal IG1, IG2 are pulse width signals whose length duration are substantially equivalent to the open times of the fuel injectors 29 and are thus representative of the amount of fuel flow. Signal IG1 is gated to four injectors beginning with a timing pulse related to RD1 and ending when the scheduled amount of fuel has been delivered. Signal IG2 is gated to the other four injectors beginning with a timing pulse related to RD2 and ending when the scheduled amount of fuel has been delivered. The cylinders are thus divided into two groups which receive fuel every engine cycle 180.degree. out of phase with the other group. To the base fuel calibration, there is added to and subtracted from special correction calibrations calculated from warm up conditions, altitude, air temperature, accelerations, cranking conditions, wide open throttle, closed throttle, exhaust gas recirculation and exhaust gas composition conditions. The water temperature sensor 35 generates the H.sub.2 O temp signal to the ECU 13 to provide the basis for a start and a warm up enrichment calibration. The water temperature sensor is located within the liquid coolant jacket of the engine 11 and provides an excellent indication of the engine operating temperature. During cold starting and initial operation of the engine, enrichment will be necessary for driveability until the engine warms to operating temperature. The H.sub.2 O temp signal is further utilized as an indication of high combustion temperatures in the cylinders. Such high temperatures will produce excess amounts of NO.sub.x which can be reduced by exhaust gas recirculation. From the H.sub.2 O temp signal the ECU 13 can sense the necessity of EGR and enable an EGR valve as will be more fully described hereinafter. Enrichment during cold cranking or starting is further aided by using the RD1, RD2 signals to sense this condition. Smoother and faster starting of the engine will be the result of the inclusion of this function. Altitude compensation providing an enrichment of the basic calibration for different altitudes is generated as a function of the MAP signal in combination with the wide open throttle signal WOT. At wide open throttle, the MAP indication will be substantially ambient pressure and thus an altitude indication. Correction to the base calibration of air flow is provided by the air temperature sensor 39 which senses the ambient air temperature coming into the intake manifold 15 and provides an electrical air temp signal to the ECU 13 which is representative thereof. This correction is necessitated because the mass air flow is less at higher temperatures than at lower temperatures for the same speed and MAP indication readings. This air temp signal is thus used to correct the air density indication of the base calibration which is the MAP signal. Pulses for acceleration and tip-in enrichment are provided from a throttle switch 41 located on the throttle body of the intake manifold 15. The throttle switch 41 senses the position of the throttle valve 23 and provides the pulsed signals AE1, AE2 to indicate the rate of change of the throttle angle or other desire for an acceleration. Further signals from the throttle switch 41 are a wide open throttle indication WOT and a closed throttle indication CTS. These signals are used as indications of these special conditions and modify other corrections as will be more fully explained hereinafter. An oxygen sensor 43 located in the exhaust manifold and sensing the oxygen content of the combustion products is provided to produce a signal 02 to the electronic control unit 13. Preferably, the oxygen sensor 43 includes a zirconia element which products a low voltage level while sensing exhaust gas compositions with an abundance of oxygen contained therein and a high level voltage signal while sensing exhaust gas compositions with an absence of oxygen contained therein. Zirconia 02 sensors have steep transitions between these levels at the stoichiometric point where the composition of the exhaust gas changes from oxygen rich to oxygen lean. An oxygen rich exhaust gas indicates not enough fuel is supplied for a stoichiometric rate which will be termed a lean air/fuel ratio and an oxygen lean exhaust gas indicates a rich air/fuel ratio where an excess of fuel has been supplied. Other input signals to the electronic control unit 13 for initialization and power include the +B signal taken from the positive battery terminal of the automobile, the ignition signal IGN from a terminal of the ignition switch 45, and the start sol signal received from the cold power terminal of the starter solenoid. As output signals, in addition to the pulse width modulated fuel control signals IG1 and IG2, the electronic control unit 13 further provides an energization signal +FP to the fuel pump 47. When the fuel pump 47 is turned on, fuel from a tank 53 flows under pressure through a filter 51 into a fuel rail 49 to which the injectors 29 are connected at one end. The pulses IG1 and IG2 then cause a solenoid operated valve in each injector to open to allow the pressurized fuel to be metered into the intake manifold 15 just ahead of the intake valve 19. Pressure in the fuel rail 49 is controlled by a fuel pressure regulator 55 which has a vacuum conduit connected to the manifold 15 for changing the regulated pressure in response to the intake manifold vacuum. The electronic control unit 13 further provides a signal +FIV to a fast idle valve 57 upon start up. The fast idle valve 57 provides an increase in the amount of air inducted into the intake manifold by opening a bypass of the throttle valve 27 during the starting conditions which will decay to a closed position after a certain passage of time. Further provided by the electronic control unit 13 is an inhibiting EGR signal, an EGR valve 59 which recirculates a portion of the exhaust gases from the exhaust manifold 25 to the intake manifold 15 in order to reduce the NO.sub.x content of the exhaust gases. As is known in the art, the EGR valve 59, when enabled, provides a fixed percentage of exhaust gas through the recirculation loop to the manifold 15 by positioning a valve with respect to the manifold vacuum. The EGR is inhibited until the engine reaches a temperature at which NO.sub.x can form. FIG. 1B illustrates a functional block diagram of the input of the sensor signals to and output of the control signals from the electronic control unit 13. Internal communication signals of the ECU 13 are further illustrated. The electronic control unit 13 is basically divided into an analog function generator 65 and a digital or microprocessor function generator 67. Analog signals air temp, H.sub.2 O temp, MAP, RD1, RD2 are input directly to the analog function generator 65. The communication of signals to the microprocessor function generator 67 include input signals WOT, CTS, and 02 that arrive directly from the engine environment and further include two internally generated signals FFS and ADS, and one of the output signals from the electronic control unit 13, the EGR signal. The microprocessor function generator 67 communicates with the analog function generator 65 by producing three output signals CLC, PDS, and LOS. The CLC signal is an analog representation of the digital output of eight signal lines via a data port 69. In the preferred embodiment the port data from port 69 is converted into an analog signal via D/A converter 71 and is used as the closed loop correction for the system. The LOS signal is utilized to give a failure indication and the PDS signal to enable a lean out function for decelerations as will be more fully described hereinafter. The division of the electronic control unit 13 into the analog function generator and microprocessor function generator serves to reduce circuitry and increase control by permitting the main analog sensor functions such as MAP, air temp and water temp to be input to an analog device which may handle them readily without analog-to-digital conversion. Digital or logic inputs that may be handled more readily by a microprocessor function generator 67 such as CTS, WOT, 02 and EGR are input directly over single lines and are either a high or low digital value. Further, the functions have been divided such that the multiplications and divisions of the electronic control unit calibrations are performed in the analog function generator 65 where they will not waste processing time or expensive memory of the microprocessor function generator. Calculations such as successive additions or subtractions, for example integrations of a closed loop correction, are performed quickly in the microprocessor function generator 67 to permit a saving of analog circuitry. The output signals IG1, IG2, EGR, +FP, +FIV which interface with analog devices are relegated to the analog function generator 65 for power amplification, and current or voltage level control. The division of functions between analog and digital sections in this manner provides an optimal system that is neither digitally oriented nor analog oriented but hybrid in nature. The hybrid design of the system substantially reduces circuitry between all digital or all analog systems and uses only one D/A converter for communication between the two function generators 65, 67. Significant cost savings are achieved by eliminating the necessity for any A/D converters. The detailed functional block diagram in FIG. 2 illustrates the combined functions of the analog function generator 65 and the microprocessor function generator 67. The generation of control signals and their communication between the various functional blocks are illustrated in the figure. The detailed circuitry for generating the described signals will be more fully discussed hereinafter. The injector control pulses IG1 and IG2 are generated by an injector driver circuit 10 from the injector drive signal IDS and the flip-flop signal FFS. The FFS signal is generated by a speed sensing circuit 16 which outputs a square wave having a positive transition for every pulse from the read input RD1 and the opposite transition for every input pulse of signal RD2. The FFS signal then is used to time the alternations between the injector groups for injection. The IDS signal is a composite signal determining the pulse width of the enabling pulse signals on the group injector lines. Usually, the IDS signal is formed by a pulse width signal PWS generated by a pulse width generation circuit 12 from the calibrations of the ECU. To the PWS signal are added an acceleration enrichment pulse signal HEP, and a closed throttle pulse signal CTP generated from an acceleration enrichment circuit. Further, the cold cranking signal CKS from the cold cranking function circuit 20 is combined to form the IDS signal. The main injector pulse width signal PWS signal is generated by a combination of signals including a speed function signal SFS from the speed sensing circuit 16, a manifold pressure function signal MFS from a pressure sensing circuit 14, and a correction current combination signal CCC from a correction current combination circuit 22. A fourth signal, reset RST, from the speed sensing circuit is provided to time the initial edge of PWS to the injector driver circuit 10. The pulse width generation circuit 12 generally begins the initiation of a pulse at the end of the reset signal at a voltage dependent on the speed function signal SFS and ending when a ramp crosses a voltage developed by the MFS signal. The ramp rate or charging speed is determined by the current signal CCC. The SFS signal from the speed sensing circuit 16 is developed by applying a functional relationship to the input signals RD1 and RD2. The speed sensing circuit 16 also generates the flip-flop signal FFS and the reset signal RST. A further signal RPM is used internally in the circuit and is generated to other circuitry in the system. The manifold pressure function signal MFS is developed by the pressure sensing circuit 14 in response to the input of a pressure sensor, the altitude compensation signal ACS from altitude compensation signal circuit 24, and the wide open throttle signal WOT. Basically, the MFS signal is a function of MAP corrected by the ACS signal and the WOT signal. The correction current combination circuit 22 combines five signals to form the CCC signal or the ramp rate of the PWS signal. The first signal, a temperature correction current signal TCC from an AB curve correction circuitry 32, is generated in response to warm up conditions and is time temperature, and load dependent. The second signal, an altitude correction current signal ACC, from the altitude compensation circuit 24 provides additional altitude enrichment in addition to that provided to the pressure sensing circuit 14 from the altitude correction ACS signal. A third signal combined in the correction current combination circuit 22 is the air temp signal from the air temperature sensor 39 which provides an analog voltage representative ambient air temperature. A further signal received by the correction current combination circuit 22 is the closed loop current signal CLC generated by the closed loop control 26. The final signal combined to yield the CCC signal is the wide open throttle signal WOT. A triangular waveform signal TWS via triangular wave generator 28 is supplied to the correction current combination circuit 22 is used to facilitate the combination of the above-mentioned five signals. A priority signal, PAE, is provided to the correction current combination 22 from the acceleration enrichment circuit 18 to inhibit the CCC signal during outputs of the acceleration enrichment pulse signal AEP and closed throttle pulse signal CTP, as will be more fully explained hereinafter. The AB curve correction circuit 32 generates a portion of the temperature correction current TCC from a time and temperature dependent warm up signal HCC developed as a function of a water temperature signal WTS and its analogy a WTS' signal from the cold cranking function circuit 20. The WTS signal and WTS' signal are generated as a function of the engine temperature from the H.sub.2 O temp signal input from water temperature sensor 35 to the cold cranking function circuit 20. Further inputs to the AB curve correction circuitry 32 are the wide open throttle signal WOT and the start signal SRT. The signal HCC is time and temperature dependent and has a delay portion generated as a signal ADS to the closed loop control 26. The HCC signal is combined with positive K correction signal generated from a positive K correction circuit 30. The positive K correction signal PKS is generated as a function of the manifold absolute pressure signal MAP via altitude compensation circuit 24 and the triangular waveform signal TWS from the triangular wave generator 28. The PKS signal and HCC signal are combined to generate the temperature correction current signal TCC. Since the MAP signal is an indication of load, the TCC signal is a warm up enrichment that is load dependent. The closed loop control circuit 26 takes the digital input signals WOT, O.sub.2, CTS, FFS and ADS to provide the closed loop current signal CLC to the correction current combination circuit 22. Further in response to the input of the EGR, ADS and FFS signals, the enabling signal PDS is generated to a deceleration lean out circuit 42. The lean out circuit 42 will generate a deceleration lean signal DLS in response to the input of the MAP signal and PDS signal. A lamp on signal LOS is generated to a failure detect circuitry circuit 38 from the closed loop control 26 in response to the FFS, ADS, WOT, EGR, and O.sub.2 signals. A fuel pump control and safety circuit 34 is provided to generate the +FP signal to the fuel pump and the +FlV signal to the fast idle valve in response to starting conditions sensed by the IGN signal. The battery voltage +B is transferred through circuit 34 to a voltage regulator where the source voltage +A is generated. A first pulse inhibit signal FPI is generated by the circuit 34 in response to the IGN signal and the start signal SRT is provided when the start solenoid is engaged and generates the incoming signal start sol. The speed sensing circuit 16 will now be discussed in further detail with reference to FIG. 3. Input signals RD1, RD2 for the circuitry are provided through terminals 100, 101 from the reed pickups of the speed sensor and transmitted via input resistors R301, R303 to inputs of NOR gates 102 and 104 respectively. A resistor R300 is connected between terminal 100 and ground to form a ground return at the input terminal. Likewise, terminal 101 has a resistor R302 connected between the terminal and ground for a ground return of the RD2 signal. A capacitor C300 connected between the input terminal of NOR gate 102 and ground filters high frequency signals from that input and a similar attenuation capacitor C301 is connected between the input of NOR gate 104 and ground. Signals RD1, RD2 are shown in FIG. 3b as alternating pulses 360.degree. of engine rotation apart and offset a set number of degrees from a timing point, for example TDC of cylinder 1. NOR gates 102 and 104 are cross-connected as to form an RS flip-flop with their output terminals feeding the input terminals of a NOR gate 106 via capacitors C302 and C303 respectively. The output of NOR gate 102 will go low or be reset when a positive voltage pulse is provided at terminal 100 and will go high or be set upon a reed pulse of signal RD2 entering terminal 101. This action will produce a square wave signal from the output of NOR gate 102 to the NOR gate 106. The output of NOR gate 104 will provide the inverted form of the square wave signal to the input of NOR gate 106. Thus, a square wave which has a frequency equivalent to the engine speed is generated at the output of NOR gate 104 and becomes the timing signal FFS output from terminal 112. Signal FFS is illustrated in FIG. 3d in timed relationship to RD1, RD2. The positive going edges of the square wave output of NOR gate 102 are differentiated by the capacitor C302 connected to the input of NOR gate 106 and a resistor R304 connected to ground to provide a negative going pulse to PNP transistor Q300 via an output resistor R306. Likewise, a differentiator comprised of the capacitor C303 connected to the input of NOR gate 106 and a resistor 305 connected to ground supplies a positive pulse upon each positive going edge of the output of NOR gate 104. This pulse is likewise used to turn on transistor Q300 via resistor R306 from the output of NOR gate 106. The differentiator values are chosen such that for each edge from the flip-flop a set duration pulse is generated from gate 106. Transistor Q300 is normally biased off by a resistor R307 connected between its base and voltage source +A, but will conduct when NOR gate 106 sinks current when in a low state. The output of the collector of transistor Q300, therefore, is a positive pulse of set duration that is developed every time one of the reed signals RD1, RD2 is present. This signal is the reset signal RST illustrated in FIG. 3e and is used as one of the main timing signals for engine speed. The RST signal is delayed for its duration in NOR gate 108 to provide an RPM signal at terminal 112. NOR gate 108 has its inputs tied together and is further connected to a pull up resistor R308 which keeps the output generally low. When, however, Q300 is pulsed, the inputs are momentarily connected to the collector terminal via a capacitor C304 at the trailing edge of the RST pulse. The resulting positive pulse is transmitted through a diode C300 to a parallel circuit combination comprising resistor R309 and capacitor C305 connected between the cathode of the diode CR300 and ground. The capacitor C305 charges to the voltage of the delayed reset pulses from the output of the NOR gate 108 during their presence and then decays to provide a voltage proportional to the engine speed at the inverting input of a differential amplifier 114. The decay is at the time constant of capacitor C305 and resistor R310. The voltage to which the capacitor decays is representative of engine speed and is the RPM signal. The RPM signal is output to various other circuits via terminal 115 and is illustrated in FIG. 3f. The amplifier 114 is configured as a comparator and has a threshold voltage connected to its noninverting input. The threshold voltage is developed at the junction of a divider resistor R311 and a divider resistor R310 connected between a positive source +V and ground. The amplifier 114 further has a positive feedback resistor R312 connected between its output and its noninverting input and includes pull up resistor R313 for the open collector amplifier connected between the positive source +A and its output. When the RPM signal voltage on capacitor C305 exceeds the threshold value that the divider resistors input to amplifier 114, its output will go low thereby sinking current away from the anode of a diode CR301 and lowering the voltage at a node 116 via an output resistor R314. Generally, node 116 is at a higher voltage which is the junction voltage of a divider resistor R316 and divider resistor R315 connected between the positive supply +A and ground. This voltage is summed with the voltage developed from a current supplied through the series combination of the pull up resistor R313, the diode CR301 and the output resistor R314 when the amplifier is off. However, the node 116 will be pulled to the lower voltage when the RPM signal exceeds the threshold of the amplifier 114. This step in voltage is thereafter transmitted to terminal 118 via a resistor 311. A capacitor C307 connected between the terminal 118 and ground provides a decay from the higher voltage level to the lower level when the RPM of the engine changes enough to switch the amplifier. This output from terminal 118 is the speed function signal SFS which is illustrated in FIG. 3g. It is seen that for RPM signal levels below 850 RPM (threshold level) the higher voltage will provide an idle lean out level which will shorten the injector pulse width and for RPM signal above a normal operation calibration level will be provided to lengthen the pulse width. The decay time constant produces a smooth transition between these two levels. It is evident the SFS signal could be a much more complex function of speed than that illustrated but it has been demonstrated that this two step function is quite advantageous and is preferred in the present implementation. The pulse width generation circuitry will now be more fully described with reference to FIGS. 4a-e. Initially, a main timing capacitor C308 connected between a node 320 and ground charges with a linear ramp toward the supply voltage because of the CCC signal sourcing a calculated amount of current thereto from terminal 322. The rate at which the capacitor charges, and thus the slope of the ramp, is determined by the amount of current supplied by the signal CCC and, as will be more fully explained hereinafter, varies with the parameters supplied to the correction current combination circuit 22. The generation of the CCC signal current and its calculation will be described with reference to the detailed description of that circuit. The voltage to which the capacitor can charge is limited by a clamp formed with a diode CR104 connected between the node 320 and the junction 321 of a divider formed with a resistor R116 and a resistor R117 connected between the source of positive voltage +A and ground. The capacitor will charge to the divider voltage plus one diode drop and remain there until the pulse width generation cycle begins. In FIG. 4d, where the voltage as a function of time for capacitor C308 is shown, this clamping voltage is illustrated and the level designated V(CLAMP). The pulse width generation cycle begins with the positive going edge of a reset pulse from signal RST of FIG. 4b turning on a transistor Q301 by energizing the base of the transistor via a resistor 319 through terminal 110. The positive going edge turns on the transistor Q301 to allow the capacitor C308 to discharge through the output terminal of an amplifier 326 to the voltage applied on the noninverting input. The voltage of capacitor 320 is applied to the inverting input of the amplifier 326 and it will equalize the voltages on its inputs during the on time of the transistor Q301. The voltage supplied to terminal 118 which is connected to the noninverting input of amplifier 326 is the speed function signal SFS and this sets the initial voltage point of capacitor 320 to its value. This voltage level is labeled as SFS in FIG. 4d and is held for the duration of the RST pulse. The positive going edge of the reset pulse RST additionally energizes a transistor Q1 to generate the inverse of the reset signal RST as shown in FIG. 4c. The base of transistor Q1 is connected to a junction of a divider comprising a divider resistor R4 and a divider resistor R5 connected between the reset terminal 110 and ground. The RST signal developed at the collector of the transistor Q1 inhibits the pulse width signal PWS generated at the output of an amplifier 330 through a diode CR7 and terminal 332 until the RST signal goes low. The RST signal is further transmitted to other circuits via terminal 334. At the falling edge of the reset pulse, the transistor Q1 and transistor Q301 are turned off. The capacitor C308 will now start charging from the CCC signal toward the clamping voltage as seen by ramp 323 in FIG. 4d. A comparator amplifier 330 compares the rising ramp voltage of the capacitor C308 at its inverting input to the MFS signal applied to its noninverting input. The voltage applied to the inverting input of the amplifier 330 as was stated is initially the speed function signal SFS via the capacitor C308. The output of the amplifier 330 is ungrounded when transistor Q1 is turned off and has a positive voltage output to the terminal 332 via a diode CR302. This is shown as point P.sub.1 in FIG. 4e as is the rising edge of the PWS signal. The capacitor 308 charges at the rate of the current source CCC and when the ramp voltage 323 exceeds the MFS voltage at the noninverting input of the amplifier 330, the amplifier will switch into nonconduction and terminate the pulse width signal PWS at point P.sub.2. Thus, it can be seen that the PWS signal is a pulse whose on time is a function of the speed function signal SFS; the manifold pressure function signal MFS; and the correction combination current signal CCC. The PWS signal is generated at every reset pulse and will be gated to the injectors by the injector driver circuitry as will be more fully described hereinafter. It is evident that the length of the pulses of the PWS signal can be shortened or extended by changing either one, two, or all three of the variable signals, SFS, MFS, and CCC. For example, raising the SFS signal will shorten the pulse width because the capacitor C308 will begin the cycle at a higher point and will have as far to charge and lowering the SFS signal will provide the opposite result. Lowering the MFS signal will shorten the pulse width and raising it will lengthen the pulse width. FIGS. 4d, 4e illustrate the effect of lowering the charging current 325, 327 which will extend the PWS pulse to P.sub.3, P.sub.4 respectively. Therefore, the pulse duration of the PWS signal and hence the amount of fuel delivered from the injectors is directly proportional to the MFS voltage and inversely proportional to the SFS voltage and CCC current. FIG. 5 illustrates the connections of the manifold pressure sensor 500 and the pressure sensing circuit 14. The MAP sensor has its differential inputs -1N, +1N, connected to ground and the source of positive voltage, +A, respectively. The +A terminal is provided with a filter capacitor C503 connected between the terminal and ground. The differential output +OUT generates a positive voltage which is an analog representation of the physical pressure detected in the intake manifold 15 of the engine. The differential output -OUT is connected to ground. The sensor may comprise a pressure bellows which changes the magnetic coupling of a differential transformer by moving a core attached thereto. A preferred sensor of this type is a linear differential pressure transducer manufactured by Gulton Industries of Costa Mesa, CA. The output of the sensor is the signal MAP which is provided to other portions of the system via terminal 502. An offset level for the MAP signal, a voltage versus pressure waveform is provided at the junction 503 of a pair of divider resistors R502, R504 connected between the source of positive voltage +A and ground. The junction 503 is connected to the +OUT terminal of the pressure sensor 500 via a trim resistor R507. The offset level and the trim resistor R507 can be adjusted so that different physical sensors of this type may be adjusted in various systems to provide identical MAP signal voltage versus pressure waveform. The offset level provides a zero adjust for an initial pressure setting and the resistor R507 is a slope multiplier. The circuit normalizes the outputs of the different sensors so the remaining system calibration does not have to be changed for each sensor. Filtering and decoupling compensation is provided by a capacitor C502 connected between the node 503 and ground. The MAP signal from the pressure sensor 500 also feeds the noninverting input of a voltage amplifier 504 having a parallel feedback loop consisting of a filter capacitor C501 and a resistor R501 connected between its output and inverting input. The amplifier 504 is connected as a noninverting voltage amplifier having a gain dependent upon the ratio of the resistance to ground. The capacitor C501 attenuates high frequency noise. The gain of the amplifier 504 is changed by two breakpoint amplifiers 506, 508 that increase the gain of the amplifier 504 by lowering the effective resistance between the inverting input and ground at certain values of the MAP signal. Initially at low MAP signal values, the amplifiers 506 and 508 are both in the non-conducting condition and block current flow through two current paths to ground. A first current path for amplifier 506 is the series connection of a resistor R503, a diode CR500 and the output of the amplifier to ground. A second current path for amplifier 508 is the series connection of a resistor R509, a diode CR503, a resistor R508, and the output of the amplifier 508 to ground. Amplifier 504, when both of these current paths are blocked, has only feedback resistance R501 connected between its output and noninverting input and, therefore, has substantially a unity gain. The output of the amplifier 504 will follow the voltage applied to the noninverting input which at these low MAP levels will be the pressure sensor output or MAP signal. This voltage is output via terminal 336 as the manifold absolute pressure function signal, MFS, via an output resistor R500. The resistor R500 further forms a low pass filter with a capacitor C500 connected between the terminal 336 and ground. The first breakpoint of the amplifier 504 at which the gain will increase occurs at a threshold voltage set on the noninverting input of amplifier 506. The threshold voltage is developed at the junction of a pair of divider resistors R506 and R505 connected between the source of positive voltage +A and ground. As soon as the voltage on the inverting input of the amplifier 504 exceeds this threshold, the inverting input of amplifier 506 will exceed it because of its connection to that point through a resistor R503. The amplifier 506 will as a consequence begin conducting thereby pulling current through the first current path and increasing the gain of the amplifier 504 by a factor. The amount of increased slope or gain is determined by the value of the resistor R503 and the breakpoint is determined by the threshold voltage. Moreover and similarly in operation, when the voltage on the inverting input of the amplifier 504 reaches the second breakpoint voltage which is set by the voltage signal ACS input through terminal 510 to the noninverting input of amplifier 508, the amplifier will conduct and pull current through the second current path which is the serial combination R509, CR503 and R508 to ground. The parallel combination of these two current paths will reduce the resistance seen by the feedback loop of the amplifier 504 even more and thereby raise the gain to a higher slope. The second breakpoint is set by the voltage supplied to the MAP sensing circuit by the altitude compensation signal ACS. This compensation signal provides one of four different voltage levels for operation of the vehicle at differing altitudes. The ACS will thus move the second breakpoint to different positions on the MFS curve as a function of altitude as will be more fully explained hereinafter with reference to the detailed description of the altitude combination circuitry. A special condition exists when the wide open throttle signal WOT, input through terminal 512 to the common anode connection of diodes CR501, CR502, goes to a high voltage. The WOT signal is fed to the noninverting input of the amplifier 506 via the diode CR501 to raise the threshold voltage at that terminal significantly above common MAP signal voltages. This will cause the amplifier 506 to be nonconductive at all times when the WOT signal is present. A high WOT signal further provides a reverse bias voltage to the junction of the diode CR503 and the resistor R508 via the diode CR502. This reverse voltage will block the conduction of current through the second current path. By deenergizing the first current path and blocking the second, the WOT signal will cause the MFS signal to follow the MAP signal as the gain of amplifier 504 will again be one. Enrichment for this condition will be supplied by a different portion of the circuit as will be more fully explained hereinafter with respect to the correction current combination circuit. FIG. 5b illustrates the graphical representation of the MFS signal voltage versus manifold absolute pressure at a reference altitude such as sea level. The reference altitude will produce the highest ACS signal voltage. The first breakpoint 514, approximately 275 torr begins that portion of the graph which is representative of normal operating conditions where fuel flow is a linearly increasing function of MAP. At the second breakpoint 516, at around 600 torr, the engine will enter a power enrichment portion of the graph where an increased fuel slope is necessitated. These increased slope conditions are preferably a partial throttle response of the engine to an increase in load. At MAP signals below 275 torr, the engine is usually in deceleration and should have the least slope of the curve. At WOT, the breakpoints are negated and the MFS signal will substantially follow the MAP signal as indicated by the linear relation labeled WOT in FIG. 5b. As was true for the SFS signal, the MFS could be a more complex function of manifold absolute pressure. However, it has been found that the double breakpoint waveform for partial throttle and the linear relation for wide open throttle is preferable and advantageous in the present implementation. In FIG. 6, detailed circuitry illustrating the voltage regulation circuit 36 and the injection driver circuit 10 is shown. The voltage regulation function is accomplished by one part of an integrated circuit 606 encapsulated in a DIP package. The integrated circuit 606 receives unregulated battery voltage +B via a terminal 600 which is connected to the terminal of the same designation on the IC package. The +B signal is also communicated to the collector of an NPN regulating transistor Q303 whose emitter is connected to a regulated power line 605. By controlling the conductance of the transistor Q303, the unregulated +B signal becomes a regulated source of voltage +A which is input to the terminal of the same designation in the IC. The IC 606 senses the +A signal and compares it to an internal reference signal to regulate the conductance of the transistor Q303 via connection at its base to the terminal CB. A capacitor C600 is connected between the base of transistor Q303 and ground for filtering purposes of the regulation control signal. The transistor Q303, therefore, acts as a series pass regulator for producing the supply voltage +A for the rest of the system. The injector drive signal IDS, which is transmitted to the TP input of the integrated circuit 606 from the terminal 332, is used to provide pulse width information to the driver circuitry. The FFS signal input to the FF pin of the integrated circuit 606 via terminal 112 is the injector timing signal used to gate the IDS signal. Driver lines 612 and 614 connected to pins D1 and D2 respectively carry the IDS signals to a set of amplifiers 616 and 618 which drive the groups of injectors via the IG1 and IG2 signals from terminal 608 and 610 respectively. The driver signals from the terminals D1 and D2 are provided by alternating or gating the IDs signals to the driver lines 612 and 614. For example, on the positive going edge and during the on time of the FFS signal, the IDS signal will be gated via input driver 614 to the driver circuit 616. On the negative edge and during the off time, the IDS signal will gate via input driver line 612 to the driver circuit 618. Current regulation during the opening and closing portions of the pulses of the IDS signal are controlled with a current driver line 620 connected through a sense resistor R607 to the +B signal. The integrated circuit package controls the current to the driver line 620 by monitoring the voltage drop across the resistor R607 via two shunt paths on either side of the resistor. The first shunt path is through the serial combination of a diode CR600 and a resistor R606 connected from the +B terminal to ground. The second shunt path is through a diode CR601 and a resistor R605 connected between the power line 620 and ground. The junction voltage of the diode CR600 and resistor R606 is communciated to the RA pin of the integrated circuit via resistor 604 and the junction voltage of the diode CR601 and resistor R605 is communicated to the CS pin of the integrated circuit. Frequency compensation and filtering are provided by a serial combination of a resistor R608 and a capacitor C602 connected between the CC terminal of the IC606 and the +B terminal. The operation and the detailed description of the integrated circuit 606 is more fully described in a copending application Ser. No. 370,140, filed on June 14, 1973 in the name of Junuthula N. Reddy and commonly assigned with the present disclosure. The disclosure of Reddy is hereby expressly incorporated by reference herein. Since driver amplifier 616 and driver amplifier 618 contain identical circuitry, only the driver amplifier 616 will be explained in more detail. The operation of the amplifier 618 will then be understood by reference to similar circuit members described in the example. Amplifier 616 is a three-stage emitter follower amplifier having a first drive transistor Q42 which is turned on by the positive going edge of the IDS pulse. The transistor Q42 has its collector terminal connected to the current line 620 via a resistor R611 and its emitter connected to the injector drive terminal 610. The transistor Q42 further has between its base and emitter terminal connected a base resistor R609 and between the emitter terminal and ground a serial combination of a resistor R610 and a capacitor C603. Turning on Q42 will energize a second stage transistor Q43 which is generally biased off through the resistor R611. The transistor Q43 has its emitter connected to the current drive line 620 and its collector connected to the injector drive terminals 610 via a resistor R612. Further connecting the collector and base is a feedback capacitor C604. The resistor R612 provides base voltage drive to an output transistor Q44 which has its collector connected to the current drive line 620 and its emitter connected to the injector terminal 610. Thus, turning transistor Q42 on will cause transistor Q43 to conduct which in turn will turn transistor Q44 on. Similarly, when the IDS signal goes low, the injector drive line will follow as each stage of the amplifier shuts off. A serial combination of a diode CR604 and a Zener diode CR602 is connected between the base of the output transistor Q44 and ground to provide a means for dissipating the flyback energy of the coils of the solenoid injectors. Since there are four injectors in each group connected to a common drive point, significant amounts of energy are stored in the magnetic fields of the coil inductances and when transistor Q44 shuts off, the voltage at terminal 610 begins to rise very quickly in the negative direction. The base of the transistor Q44 being turned off by the transistor 43 will, however, begin to follow this voltage until the base of the transistor Q44 becomes negative enough to turn the Zener diode on through diode CR604 and it thus will clamp at its Zener voltage. The energy may then be dissipated through the emitter-collector junction of the transistor Q44 instead of the base-emitter junction. The energy can be dissipated very quickly and easily in this manner and provides an advantageous benefit in that the gain of the transistor can be used. By dissipating the energy through the transistor collector to emitter path, the wattage rating of the Zener diode may be reduced by the gain of the transistor. Illustratively, if a 30-watt diode was necessitated to dissipate the energy from the four injector coils before, a transistor with a gain of 30 would now allow a 1-watt Zener diode to be utilized thereby producing a significant cost savings. FIG. 7 illustrates the detailed circuitry of the correction current combination circuit which combines the various correction current signals TCC, ACC, CLC, air temp, and WOT to provide the slope or ramp rate of the charging current to the timing capacitor C308 of the pulse width circuit. The current signal CCC is generated as a charging current at terminal 180 through the serial path of a load resistor R19 and the emitter-collector junction of a current source transistor Q22. The base of the current source transistor Q22 is connected to the base of a mirror transistor Q21 which has its emitter connected to the source of positive voltage +A through a load resistor R118 and its collector connected to the cathode of a diode CR103 at a voltage node 178. The commonly connected bases of both transistors Q21 and Q22 are further connected by the diode CR103 to the voltage node 178. By controlling current pulled through the mirror transistor Q21 by a control transistor Q20 and hence the voltage at node 178, the current supplied through the transistor Q22 can be accurately controlled to mirror this value. The greater the current supplied through the transistor Q22 to terminal 180 the shorter will be the resulting injector pulse width because of the faster charging of the timing capacitor of the pulse width generation circuit and conversely supplying less current will extend the pulse duration. The amount of current drawn by the control transistor Q20 and the voltage at node 178 is dependent upon and controlled by a voltage controlled current sink circuit 175. The voltage controlled circuit 175 includes an operational amplifier 176 having its output connected to the base terminal of the control transistor Q20. The transistor Q20 has its collector connected to the voltage node 178 and its emitter connected to one lead of a voltage resistor R114 whose other lead is connected to ground. A voltage node 174 which is the junction of the emitter of the transistor Q20 and the resistor R114 is further connected to the inverting input of the amplifier 176. The noninverting input to the amplifier 176 is connected to a voltage node 172 which controls the voltage at the node 174 by causing the control transistor Q20 to draw enough current through the transistor Q21 to equalize the voltages between the node 172 and 174 or the inputs of the amplifier 176. Thus, the current signal CCC can be controlled by varying the voltage at node 172. The voltage at the node 172 is developed across a current summing resistor R111 connected between the node and ground. The current supplied to the resistor R111 is generated from combining the output of two devices; the first from a multiplier amplifier 164 and the second from a current source amplifier 170. The current source amplifier 170 supplies current to the node 172 and the multiplier amplifier sinks current away from the node 172. These devices will now be discussed separately with the current source amplifier 170 the initial point of description. Current that is transmitted by the amplifier 170 to the node 172 is controlled by a driver transistor Q19. The driver transistor Q19 is connected at its emitter to the source of positive voltage +A through a load resistor R113 and further connected at its collector to the node 172. Control is provided to the transistor Q19 by connecting the output of the amplifier 170 to the base of the transistor and feeding back a voltage signal to the inverting input of the amplifier from the emitter of the transistor. The amplifier has a bias voltage connected at its noninverting input which is the junction voltage at node 167 of a pair of divider resistors R108 and R112 connected between the source of positive voltage +A and ground. The bias voltage then causes the amplifier 170 to try to equalize the voltage at its inputs and thereby controls the current sourced through the transistor Q19 and the voltage at node 172 as a function of the voltage input to the threshold node 167. A quiescent current flows through the summing resistor R111 from the transistor Q19 as a result of the bias voltage at node 172. Along with a quiescent current from the multiplier amplifier 164, this current will set a base calibration voltage on the node 172 and consequently a base calibration current to flow from the terminal 180. At this point, the ramp of the charging capacitor C308 will be set and the system will deliver a pulse width as a function of only the MFS and SFS signals as previously discussed. By varying the current signal, CCC and thus the slope of the charging ramp as a function of other variables of the engine the pulse width to the injectors can be corrected for a number of physical operating conditions. The first correction to the base RPM, MAP calibration is for an air temperature calibration and is applied to the node 167. The voltage at the node 167 is increased by an inverting amplifier 168 which receives the air temp signal via terminal 166 and delivers a positive current to the node 167 from its output. The output of the amplifier 168 is proportional to the air temperature and varies the drive of the amplifier 170 at the node 167 in correspondence with this signal. This will cause less current to be supplied to the node 172 thereby lengthening the pulse width and providing greater enrichment for cold air temperatures. This, of course, is needed because the air flow is becoming increasingly denser as the ambient temperature decreases. As the air temperatures become more elevated, a lean out is provided to shorten the pulse widths. The circuitry comprising the inverting amplifier 168 is formed by providing a positive bias voltage at the noninverting input of amplifier 168 which is the junction voltage of a pair of divider resistors R101 and R109 connected between the source of positive voltage +A and ground. The input to the inverting input of the amplifier 168 via an input resistor R105 is developed at the junction of the combination of a resistor R106 and the air temperature sensor connected between the source of positive voltage +A and ground. A negative gain for the amplifier 168 is generated by connecting a feedback resistor R107 between the output, which is developed through a blocking diode CR102, and the inverting input. The blocking diode CR102 permits the amplifier 168 to source current to the node 167 but not to sink current from it. In operation, as the air temperature increases and the resistance of the air temperature sensor goes up, the voltage at the inverting input of the amplifier 168 will increase causing the voltage output at the node 167 to decrease. This increasing voltage at the inverting input will shorten the pulse width by increasing the current output from terminal 180 proportionately. At this point, the speed density calibration of the ECU is complete as the SFS and MFS calibration has been corrected for air density variations on account of air temperature. The effect of the air temperature correction on pulse width is illustrated in FIG. 7b where the enrichment is greater at lower temperatures than at higher temperatures. An incremental change in temperature at the sensor will cause a linear incremental change in the current from transistor source Q19. Since the voltage at node 172 and hence the pulse width will change as 1/R for an incremental changes in current, the enrichment will vary as 1/T as illustrated in the figure. The closed loop correction signal CLC may also be connected to the emitter of the transistor Q19 to provide a closed loop current correction to the pulse width. The closed loop signal is joined into the circuit at this point to allow the basic correction to be changed by a closed loop correction based upon the signals from an exhaust gas composition sensor. Closed loop circuits for providing an integral control signal based upon the oxygen content of the exhaust gas are conventional in the art. Since a closed loop signal of this type is generally clamped during start and warm up conditions, the signal should not adversely affect those calibrations and, therefore, is isolated from the start and warm up circuitry in this manner. In the referenced system, closed loop control is provided by regulating a current sink to vary the amount of current applied to the node 172. Normally, when the system is operating under open loop control the current sink is operable to draw a fixed amount of current from the emitter of Q19 to form a midpoint value. The current sink is subsequently regulated under closed loop control to either lengthen the pulse width by drawing more current away from emitter Q19 or to shorten the pulse width by drawing less current away from the emitter of the source transistor. If the closed loop control is used with the system, the quiescent value of current to the resistor R111 will take into account the midpoint value for open loop system operation. The output of the multiplier amplifier 164 is connected to the current summing resistor R111 via a resistor R110. The amplifier further has a threshold voltage connected to its inverting input which is the junction voltage of a pair of divider resistors R100 and R103 connected between the source of positive voltage +A and ground. An integrating capacitor C101 is further connected between the inverting input of the amplifier 164 and ground. The noninverting input of the amplifier 164 receives the TWS signal from the triangular waveform generator via terminal 158. On the upward ramp of the triangular waveform TWS the noninverting input of amplifier 164 will exceed the voltage set at the inverting input and the output of the amplifier will transition to a high level. The triangular waveform will then peak at a voltage of +6 V and begin a downward ramp and, as that ramp voltage crosses the inverting input voltage, will cause the output of the amplifier 164 to transition to a low level. In this way, the output of the amplifier 164 will be a squarewave whose duty cycle is dependent upon the voltage at the inverting input of the amplifier 164. The threshold voltage developed by the resistors R100 and R103 is chosen to form a quiescent duty cycle which, when output through the resistor R110 and summed in the resistor R111 at node 172 will be just below the TWS signal. An integrating capacitor C106 connected between the node 172 and ground produces the voltage component from the multiplier amplifier 164 as a linear function of the duty cycle of the pulses. The TWS signal is illustrated in FIG. 7c as a triangular waveform that ramps between a positive voltage value and substantially ground level. Any voltage over the threshold will produce positive pulses from the output of the amplifier 164. The duration of the positive pulses may be decreased by increasing the voltage at the node 161 which will thereafter decrease the voltage at the node 172 by sinking more current and thus reduce the charging current from terminal 180 to lengthen the pulse width signal to the injectors and provide enrichment. The warm up current signal TCC enters node 161 via the terminal 162 and diode CR100 to produce an increase in the voltage on the resistor 103 proportional thereto. The greater the warm up current, the greater the conducting time of the output duty cycle of the amplifier 164 will be and thus the smaller the current that will be delivered from the CCC signal. The result of an increasing warm up signal TCC will be to lengthen the pulse width for warm up enrichment. The TCC signal is in the preferred embodiment a variable amplitude current with a variable duty cycle. As will be more fully explained hereinafter, the duty cycle of the "on" time of the TCC signal will be increased with the MAP signal to provide increased enrichment for heavy loading of the engine while the amplitude of the signal will be varied with time and temperature. The TCC signal is illustrated in FIG. 7c as a voltage level that varies above the threshold level of amplifier 164 to decrease the pulse width to a value T.sub.1 in FIG. 7d and to a value T.sub.3 in FIG. 7e. Further enrichment is provided by providing the WOT signal to be summed at the node 161 via a resistor R102 and a diode CR100 through terminal 160. A capacitor C100 is connected between the resistor and diode junction and ground to provide low pass filtering for the basically digital WOT signal. Resistor R102 is chosen as a value to provide an increase in enrichment voltage at the node 161 in addition to the threshold voltage or the voltage developed by the TCC current signal. This extra voltage from the WOT signal will cause the shortening of the duty cycle of the amplifier 164 and thus a reduction in the CCC current output and a lengthening of the pulse width. The WOT enrichment is needed for engine power at wide open throttle and overrides the base calibration. Voltage level WOT as illustrated in FIG. 7c causes a shortening of the pulse width from the amplifier 164 to a value T.sub.2 as seen in FIG. 7e. Beyond the speed-density calibration of the ECU 13 and correction currents of FIG. 7, there is an additional correction factor for air/fuel ratio during the special condition of start up. Although the start up period is relatively short in duration, correct air/fuel ratio control during this period is critical for driveability. The cold cranking function circuitry 20 is utilized for an enrichment of the air/fuel ratio during start up and supplies pulse width information to the injector driver directly via the CKS signal. The cold cranking function circuitry will now be more fully described with reference to FIGS. 8a-e. The cold cranking function circuit provides extra temperature dependent fuel pulses to enrich the air/fuel ratio during cranking for quick and easy starting of a cold engine. The circuit comprises a comparator 204 which has an output connected to a terminal 206 via a diode CR201. Terminal 206 provides the cold cranking pulses as a signal CKS to be ORed with the main fuel pulses. These pulses will usually be larger than the PWS signals and thus overlap them. When the engine starter solenoid is disengaged, the CKS signal will be inhibited and the PWS signal will then supply the scheduled fuel to produce a smooth transition from the starting condition. One input to the amplifier 204 via the inverting input is supplied via the junction voltage 205 of a divider combination consisting of a resistor R210 and a resistor R212 connected between a supply voltage node 203 and ground. The supply voltage node 203 is essentially at the positive voltage supply +A and is decoupled and filtered during cranking of the engine starter when the voltage regulation of the vehicle is somewhat irregular by a decoupling resistor R209 and a filter capacitor C202. The junction of the divider, node 205, is periodically grounded through the collector-emitter of a transistor Q17 and a timing resistor R211. The emitter of the transistor Q17 is connected to ground and its base is connected via a resistor 218 to terminal 110. Further connected at the collector of the transistor Q17 is one terminal of capacitor C203 which has its other terminal connected to ground. Terminal 110 is a source of the reset signal RST illustrated in FIG. 8b. The RST signal grounds node 205 by causing the transistor Q17 to conduct on the positive going edge of the pulse and subsequently holds the node at ground for the duration of the pulse thereby discharging the capacitor C203. At the termination or the falling edge of the reset pulse, the transistor Q17 will turn off and capacitor C203 will charge exponentially to the divider voltage of node 205 set by the resistors R210 and R212. The charging time for the increasing voltage will be the RC time constant of the capacitor C203 and the resistor R211. The voltage waveform at node 205 is illustrated in FIG. 8c as V(N205). A variable threshold voltage is supplied to the noninverting input of the amplifier 204 at node 201 via an input resistor R214 from a water temperature sensor circuit 211. This threshold voltage varies with the temperature of the coolant of the engine and is an indication of the operating temperature of the engine. Decreasing voltages at node 201 indicate increasing engine temperature. The amplifier 204 will then switch between conduction and nonconduction to generate the cold cranking pulses by comparing the voltages at nodes 201, 205 that are applied to its inputs. The variable threshold voltage VN201 is illustrated in FIG. 8c. Cold cranking pulses will begin at 220 in FIG. 8d on the initial edge of the reset pulse when node 205 is grounded and drops the inverting input voltage below the threshold voltage supplied to the noninverting input. The pulse will terminate at 222 when the voltage at node 205 exceeds that of node 201 at 224. The pulse length is dependent upon the level of the threshold voltage and the timing constant of the capacitor C203. Colder temperatures or higher threshold voltages will cause the exponentially increasing voltage on node 205 to cross the threshold later in time than when the engine is warmer and thereby increase the pulse width as seen by pulse 226. Lower threshold voltages developed at higher engine temperatures will cause shorter pulse widths such as pulse 228. A positive feedback loop for the amplifier 204 is provided by the series combination of a resistor R215 and a diode CR200 connected between the output and noninverting input. The circuit is further provided with an active pull up for the amplifier 204 via the resistor R235 connected between the output of the amplifier 204 and terminal 212. The SRT signal is received at the terminal 212 and enables the CKS signal only during its high voltage state as seen in FIG. 8e. Thus, the CKS signal will be inhibited when the start signal is not present i.e. when the starter solenoid releases. The operation of the amplifier 204 is additionally inhibited by the amplifier 208 having a threshold voltage at a junction 213 of a pair of divider resistors R217 and R216 connected between the source of positive voltage +A and ground. The inverting input of the amplifier 208 receives the RPM signal, a voltage proportional to the RPM of the engine, via terminal 115 and compares it to the threshold voltage. The amplifier 208 is an open collector type that grounds the base of the transistor Q17 through its output terminal once the RPM signal voltage exceeds the threshold. Preferably the threshold voltage is representative of approximately 325 RPM which minimizes over enrichment during starter overrun. The threshold is illustrated in FIG. 3f as the voltage level labeled RPM(CK) or the RPM cranking threshold. The temperature dependent threshold voltage at the node 201 is developed by a noninverting voltage amplifier 202. Amplifier 202 is connected at its output terminal to the base of a PNP transistor Q12 that provides current drive from a pull up resistor 208 connected to the source of positive supply +A. The current is used to develop a voltage across a resistor R206 connected between the collector of transistor Q12 and ground. The gain of the amplifier 202 is determined by a feedback resistor R207 connected between the inverting input and the emitter terminal of transistor Q12, the parallel combination of a resistor R205 and a resistor R204. An offset of approximately one volt is provided to the inverting input by the divider combination of a resistor R204 and the resistor R205 connected between the source of positive voltage +A and ground. The H.sub.2 O signal is input to the amplifier 202 at its noninverting input is via an input resistor R203 which is fed from the junction of a resistor R207 and the water temperature sensor 35 connected between the source of positive voltage +A and ground. The water temperature sensor 35 is a variable resistance which will vary the junction voltage of the sensor and the resistor R202 between approximately 4 and 6 volts in response to changes in the coolant temperature. This voltage is proportionately amplified to provide a swing from approximately 4 to 9 volts at the emitter of transistor Q12 and thereby a proportional voltage at the junction of its collector and resistor R206. This voltage proportional to the coolant temperature is thereafter filtered by a low pass filter consisting of a resistor R213 and a shunt capacitor C200 which transmits the voltage to the input resistor R214 of the amplifier 204. The water temperature circuit 211 further provides the water temperature signal WTS and its analogy WTS' to various other parts of the system via terminals 230, 232 respectively. Signal WTS is used to sink current into the output terminal of amplifier 202 and provides an increasing conductance for decreases in engine temperature while the WTS' signal is a voltage from the emitter of Q12 which increases with temperature. The WTS, WTS' signals are utilized mainly in the warm up correction circuitry which will now be described in detail. The circuitry comprising the A and B curve warm up correction is shown to advantage in FIG. 9a. The A curve circuitry provides a warm up correction that is time and engine temperature dependent while the B curve circuitry provides a warm up correction based on engine temperature only. Each of these circuits produce an enrichment current which is summed with the other with the total then corrected for engine load as will be more fully explained during the description of the positive K circuit. The B curve current is supplied by two parallel source transistors Q13 and Q14 to the terminal 240 via their common collector connection at node 224. When combined at node 248 with the A curve current produced via a source transistor Q16, the terminal 240 generates a total temperature current signal HCC. The transistor Q13 is connected at its base to terminal 230, at its emitter to a load resistor R221, and at its collector to the current summing node 224. The other terminal of the load resistor R221 is connected to the junction of a voltage divider having a divider resistor R219 and a divider resistor 222 connected between the source of positive voltage +A and ground. The transistor Q14 is configured similarly with its base to the terminal 220, its emitter to a load resistor R223 and its collector to the current summing node 224. As was the case for the load resistor 221, the load resistor 223 is connected at its other terminal to the junction of a voltage divider having a divider resistor R220 and a divider resistor R224 connected between the source of positive voltage +A and ground. In operation, the B curve current generator receives the water temperature signal, WTS, that varies with the engine operating temperature via the terminal 220. At low temperatures, the WTS signal is substantially lower (sinks more current) than at high temperatures and therefore Q13 and Q14 are fully on. Node 224 then receives the full current available from the divider voltages through the load resistors 221 and 223. The divider voltages are different and the lowest voltage will be the first breakpoint 261 of the B curve versus temperature schedule illustrated in FIG. 9c. When the WTS signal becomes greater than the first divider voltage, the reversed biased base will shut the transistor off and provide only one current source to the summing node 224. As the signal WTS continues to rise, the second transistor will thereafter shut off at the second divider voltage shown as point 262 in FIG. 9c forming a fully warmed up schedule. Therefore, at higher engine temperatures the B curve will supply less current than at lower temperatures until the current becomes zero at a temperature equivalent to the second divider voltage. The divider voltages or breakpoints for the curve can be set at various positions but preferably the first point 261 is set at approximately 60.degree. F. and the second is set at approximately 150.degree. F. Below 60.degree. F., the increased slope of the curve provides necessary enrichment to operate the cold engine, and above 150.degree. F. the B curve enrichment is no longer required as the full engine operating temperature has been reached. The A curve current generation and enrichment schedule, illustrated in FIG. 9b, will now be discussed in greater detail. The A curve current source transistor Q16 operates in a similar manner to the B curve source transistors having its base connected to a node 250 which supplies it with a variable voltage that is time and temperature dependent and having its collector connected to the node 248 which sums the A and B current. The emitter of the transistor Q16 is connected via an emitter resistor R253 to the junction point of a divider consisting of the serial combination of resistors R248, R254 and a divider resistor R252 connected between the source of positive voltage +A and ground. A low voltage applied to the node 250 will produce a maximum current out of the transistor Q16 to the node 248. As the variable voltage which is time dependent begins to rise at the node 250, less and less current will be supplied to the node 248 through the transistor Q16. When the voltage at the node 250 surpasses the breakpoint or the divider voltage supplied to the emitter of Q16, the current source will shut off. The time dependent voltage at node 250 is generated by controlling voltage at a resistor R251 connected between the node and ground via an emitter follower. The emitter follower is formed by a transistor Q18 having its collector connected to the source of positive voltage +A and its emitter connected to the voltage node 250. The drive voltage for the emitter follower transistor Q18 is provided by the node 252 which is the collector of a transistor Q15 which is initially in a conducting state. The transistor Q15 has its emitter connected to a voltage node 254 which forms the junction of a pair of divider resistors R246 and R249 connected between the source of positive voltage +A and ground. During the time the transistor Q15 is in conduction the node 252 and node 254 are only slightly different in voltage with a capacitor C207 being fully discharged through the transistor Q15 by being connected to its collector at one terminal and to its emitter at the other. The voltage at the node 254 is dependent upon the ratio of the divider resistors R246 and R249 and upon the amount of current that is taken through the diode CR202 from the node by the WTS signal via terminal 220. As the temperature rises, less current is drawn through the diode CR202 and therefore the voltage rises on node 254 and 252, respectively, thereby driving the transistor Q18 further into conduction and the current source Q16 further out of conduction. This will supply a temperature dependent starting current for the A curve schedule as shown for the increasing temperatures T.sub.1 -T.sub.3 in FIG. 9b. The graphs illustrate that for increasing temperatures the initial currents for the A curve will decrease. However, at temperatures above T.sub.4 the WTS signal will not draw any substantial current away from node 254 and the initial A curve current will be set by the divider voltage for the figure it is seen the initial starting points on the enrichment axis are identical for T.sub.4, T.sub.5 and T.sub.6. The transistor Q15 is initially in a conducting state via a positive bias consisting of the serial combination of a bias resistor R243 and a bias resistor R245 connected between the source of positive voltage +A and the base of the transistor. Connected to the junction of these bias resistors is a pull-up resistor R261 also connected to the output of a switching amplifier 256. The amplifier 256 is normally turned off or nonconductive via a positive bias on its noninverting input. This bias voltage is developed by connecting the noninverting input to the junction of a pair of divider resistors R237 and R240 that are connected between the source of positive voltage +A and ground. The noninverting input further has a hysteresis resistor R242 connected between the output of the amplifier and the input. The inverting input of the amplifier 256 is connected to the junction of a timing resistor R238 and a timing capacitor C206 connected between the source of positive voltage +A and ground. A transistor Q11 having its collector connected to the junction of the capacitor C206, resistor R238 and having its emitter connected to ground forms a means for discharging the capacitor 206 when the transistor is turned on by the start signal SRT provided through terminal 260 via an input resistor 236. In operation, the SRT signal will cause the transistor Q11 to discharge the capacitor C206 and begin a timing cycle wherein the capacitor starts charging to the supply voltage with a timing constant determined by the resistor 238 and its capacitance. The timing cycle begins after the start signal SRT goes low which indicates the starter solenoid has opened. During the timing cycle of R238, C206 the transistor Q15 is conductive and will hold node 252 substantially at the temperature dependent voltage developed at node 254. This action generates the part of the schedule in FIG. 9b labeled hold. It is seen for different temperatures that different holding levels of enrichment current are generated at T.sub.1, T.sub.2, and T.sub.3. The hold levels are then time dependent extensions of the initial A curve currents set by the voltage at node 254. As the voltage at the inverting input of amplifier 256 passes through the threshold voltage of the divider R237, R240 the amplifier will ground the base of transistor Q15 through its output turning the transistor off. After the base of transistor Q15 is grounded, the circuit begins charging the capacitor C207 from the positive supply +A via a charging path consisting of the series combination of the resistor R247 and the resistor R248. The increasing voltage at node 252 because of capacitor C207 causes the current from transistor Q16 to decrease. Since the capacitor voltage at node 252 is exponentially increasing the current supplied at the collector of transistor Q16 will exhibit an exponential decay. The action of the amplifier 256 therefore holds the transistor Q16 into conduction for a set time period at a current dependent upon the temperature. Once the time period has elapsed, the current will decay to a fully engine fully warm value or in the preferred case decay to zero. This is illustrated in FIG. 9b and the portions of the graph labeled decay. Another input to the noninverting terminal of the amplifier 256 is from an amplifier 260 via an output pull-up resistor 241. The amplifier 260 is connected in a switching mode with a hysteresis resistor R227 connected between its output lead and the noninverting input. The noninverting input further has a threshold level applied to it from the junction of a divider consisting of a divider resistor 226 and a divider resistor 225 connected between the source of positive voltage +A and ground. The inverting input to the amplifier 260 is connected to the terminal 222 which is provided with the WTS' signal. In operation, the WTS' signal is a measure of the engine operating temperature as was the WTS signal and is an increasing voltage with temperature. Therefore, as the engine temperature increases the inverting input of the amplifier 260 will become closer to the threshold of the divider R225, R226. As it passes through the threshold value, the amplifier 260 will ground the noninverting input of amplifier 256 through the output resistor 241. This essentially defeats the holding time constant of the capacitor C206 and the resistor R238 as the inverting input will almost immediately rise above ground thereby switching the amplifier 256 on and thus turning the transistor Q15 into a nonconductive mode. The result of this hold defeating feature is illustrated in FIG. 9b at temperature T.sub.6 where there is no holding period for the A curve current schedule. The threshold of amplifier 260 corresponds to the temperature T.sub.6 and is selected such that the warm up and idle operation of the engine is not affected adversely. The holding time for current levels at any temperature above T.sub.6 is also defeated as is shown by the curve for temperature. These hold defeat features permits enrichment when necessitated for driveability during warm up but reduces emissions and improves fuel economy when the holding function can be eliminated at high engine temperatures. The operation of the EGR signal circuit will now be more fully explained. The EGR valve solenoid which is connected to the collector terminal of a transistor Q9 receives positive power voltage +B via a load resistor R255 when the transistor is turned on. The diode CR206 connected between the terminal of the EGR valve solenoid and ground provides a discharge path for the collapsing field of the solenoid coil when the transistor Q9 turns off. A pair of diodes CR204 and CR205 are serially connected between the base of the transistor Q9 and the +B terminal to limit the current flow through the transistor and hence the solenoid by providing a set voltage drop across the junction. A signal transistor Q8 will turn on the power transistor Q9 by grounding one terminal of a resistor 256 connected at its other terminal to the base of the transistor Q9. A bias resistor R257 is connected between the +B power terminal and the base of transistor Q9 to provide reverse bias to the base of Q9 when the transistor Q8 is nonconducting. Usually, the signal transistor Q8 is in a conducting state and therefore Q9 is in a conducting state. The bias for the conducting mode of Q8 is provided by a positive bias to the base terminal of the transistor. The bias is developed at the junction of a divider consisting of the serial combination of resistors R231, R234 and a resistor R258 connected between the source of positive voltage +A and ground. A switching amplifier 258 is operable to turn transistor Q8 on or off is connected at its output terminal to the junction of the resistor R231 and the resistor R234. The amplifier 258 further has a hysteresis resistor R232 connected between its output terminal and its noninverting input. The amplifier 258 is provided with a threshold voltage at its noninverting input via the junction of a divider consisting of a divider resistor 229 and a divider resistor 228 connected between the source of positive voltage +A and ground. A control input signal for the amplifier 258 is provided by the junction of a filter capacitor C204 and an input resistor R230 the combination being connected between the terminal 222 and ground. The inverting input and the noninverting input of the amplifier 258 further has a delay capacitor C205 connected between them. The WTS' signal is input to the terminal 222 as an engine temperature indication. In operation, the EGR signal is generally high engaging the EGR solenoid and positioning the EGR valve in order to block EGR flow from the engine. However, when the WTS' signal exceeds the threshold set by the divider 229, 228 at the noninverting input of the amplifier 258, the amplifier grounds the base of the transistor Q8 through its output terminal via the resistor R234. As the transistor Q8 becomes nonconductive, the transistor Q9 also turns off and the EGR valve solenoid disengages the EGR valve and supplies the engine with exhaust gas recirculation of a certain percentage. This operation provides exhaust gas recirculation only when the engine operating temperature increases to the temperature required to generate substantial quantities of NO.sub.x which is represented by the threshold of amplifier 258. During these elevated temperature conditions control of the position of the valve is a function of manifold absolute pressure (intake manifold vacuum) as has been described previously. A resistor R259 couples the WOT signal via terminal 266 to the base of the transistor Q8. When the wide open throttle signal goes high, the EGR valve will be closed even if the operating temperature of the engine is sufficient to pull Q8 out of conduction. This is a desirable operation during full power conditions indicated by the WOT signal, where exhaust gas circulation could be detrimental to power generation. This feature can be excluded if a timing adjustment is made for EGR at wide open throttle. The linear positive K circuit which provides a load dependent correction to the warm up calibration of the A and B curve currents will now be more fully described with reference to FIGS. 11a, 11b. The linear positive K circuit comprising a multiplier amplifier 310 and a voltage amplifier 308 receive an input via a terminal 309 which is the MAP signal. The MAP signal is proportional to the load of the engine and is utilized by the circuit to enrichen the air/fuel ratio during high load conditions. The MAP signal is transmitted to the inverting input of the amplifier 308 via an input resistor R132. The noninverting input of the amplifier 308 is supplied with a threshold voltage via the junction of a voltage divider comprising a divider resistor 134 and a divider resistor R133 connected between a source of positive voltage +A and ground. The output of the amplifier 308 is fed via its base terminal to a PNP transistor Q26 connected at its emitter to the inverting input of the amplifier and to one terminal of a capacitor C105 at its collector. The other terminal of the capacitor C105 is connected to ground. The transistor Q26 and amplifier 308 configuration provides a thresholding amplifier with a unity gain. The threshold level is provided as representative of an absolute pressure in the intake manifold of approximately 325 torr which is the point at which load begins to be a factor in the warm up enrichment schedule. The signal on capacitor C105 which is the MAP signal voltage if over the threshold voltage is presented at node 311 of the multiplier amplifier 310 via its noninverting input. A bias voltage supplied to node 311 is added to the capacitor signal before it is input to the amplifier 310. The bias voltage for the multiplier amplifier 310 is provided at the junction of a voltage divider comprising a pair of divider resistors R131 and R130 connected between the source of positive voltage +A and ground. The TWS signal scaled by a pair of divider resistors R128, R129 connected between terminal 307 and ground is input to the inverting input of amplifier 310 at their junction. The triangle waveform signal TWS via the inverting input of amplifier 310 is mixed proportionately with the voltage formed at node 311 to provide a variable duty cycle signal, PKS, proportional to the pressure signal MAP. The operation of multiplier amplifier 310 is similar to the multiplier in the correction current combination circuit previously described. The bias voltage at node 311 provides a base duty cycle which is varied by the MAP signal in a linear manner. This linear positive K signal PKS which is represented by a square wave with a variable duty cycle is then output via the terminal 312 to be combined with the current from terminal 240 which comprises the HCC signal. The combination of the PKS signal and the HCC signal is termed the temperature correction signal, TCC, and is input to the correction current combination circuit at terminal 160. The actual combination or linear multiplication in the present implementation is performed by sinking the HCC current into the output terminal of amplifier 310 when the TWS signal is greater than the voltage at node 311 and applying full HCC current to terminal 160 when the TWS signal is lower. Since the voltage at node 311 rises with an increasing MAP signal, greater amounts of the HCC current will be applied to terminal 160 at greater loads thereby providing more enrichment. Thus, by varying the on/off time of the PKS signal, the warm up current from the A and B curve generators can be made to vary linearly with the load of the engine. The terminal 162 is one input drive terminal for the multiplier amplifier 164 which performs a linear multiplication of its inputs and changes the pulse width to the injectors accordingly as was previously described. Therefore, by inputting the TCC current at the terminal 162, the relationship between the warm up current and MAP will not change except by the scaling factor of multiplier amplifier 164. With reference now to FIG. 11b, there is shown a three-dimensional representation of the warm up current as a function of MAP and temperature. The percentage of enrichment is the ordinate variable, with temperature represented on the X-axis and decreasing MAP forming the Z-axis variable. The heavy dark line 316 in the X-Y plane is the B curve current schedule of FIG. 9c and can be projected along the Z-axis by lines 318, 320, 322 to form a solid figure that does not vary with the PKS signal. However, if, as in the preferred embodiment, an enrichment factor that increases linearly for a rise in MAP or load is drawn at line 324, and enrichment surface bounded by lines 320, 324, 326, 328, and 330 is formed. The solid formed in such a manner is the B curve current surface after providing linear positive K to the B curve current schedule. It is evident there is quite a significant change in warm up enrichment between those areas where the need is the greatest (cold engines under high load) and those areas where the enrichment factor is the least significant (warm engines and light loads). The most enrichment change between the present system and prior systems, however, occurs at light loads and cold temperatures. The parabolic line 332 illustrates prior enrichment schedules where the greatest effect of positive K is at high loads, generally up in power operating region of the MAP schedules. It is evident by comparing the lines 324 and 332 that the shaded area of enrichment between the two has been lost by this method. The normal operating region of the MAP curve (300 torr-600 torr) is then provided with significantly more enrichment by the linear method. This schedule, it has been found, improves the driveability and response of the engine during the warm up period. Similarly, the A curve of FIG. 9b is represented by a surface bounded by lines 334, 336, 338, 340, 342, 344. Line 334 illustrates the A curve schedule at minimum values of positive K and line 336 illustrates the A curve schedule at maximum values. The A curve surface, as has been noted before, will decay into the B curve surface with respect to time. The time variable has not been illustrated in FIG. 11b for the purpose of clarity. With reference now to FIG. 10, the detailed circuitry comprising the circuitry for generating the triangular wave signal TWS of FIG. 7b will now be more fully explained. The triangular waveform generator is basically comprised of an integrating amplifier 302 and an inverting amplifier 300. Amplifier 302 has two current inputs, one is provided via a resistor R127 connected between a source of positive voltage +A and its noninverting input and the other current is provided from a series connection of a resistor R126 and a resistor R125 connected between a source of positive voltage +A and its inverting input. A capacitor C104 is connected between the inverting input and the output of the amplifier 302. Depending upon which current input (to the inverting or noninverting input) is larger, the integrating amplifier 302 will cause the output terminal 307 via the integrating capacitor to be either ramping toward +A or ramping toward ground. The input resistors R125, R126, and R127 are sized such that when node 304 is at a positive potential, the current flow into the inverting input is greater than the current flow into the noninverting input and thus the output of the amplifier 302 ramps from the source voltage +A to ground. When, however, a switching transistor Q25 pulls the node 304 to ground, the current is supplied by R127 to the noninverting input will cause the output of the amplifier 302 to ramp from ground to the source voltage +A. The current supplied by the series combination R126, R125 is twice the current supplied to the input by resistor R127 and therefore the output of the integrating amplifier 302 is symmetric and generates a triangular waveform between ground and the positive supply +A. The feedback from the output of amplifier 302 to the inverting amplifier 300 makes the oscillation stable at a set frequency. A threshold is provided to the noninverting input of the amplifier 300 via the junction of a divider resistor R120 and a divider resistor R121 connected between the positive supply +A and ground. The threshold voltage is approximately one half of the supply voltage +A. Further connected to this junction point is a resistor R122 which feeds a node 306. Attached to the node 306 is a parallel combination of a capacitor C103 and a resistor R124 generating an input signal to the base of the transistor Q25. The base of the transistor Q25 further has a divider resistor R123 connected between the base terminal and ground. The output of the amplifier 300 further feeds the node 306. Node 306 is generally held at the output voltage of the amplifier 300 and when the output voltage of integrating capacitor 302 exceeds or falls below the threshold at the noninverting input, transistor Q25 should switch into a nonconducting or conducting mode respectively. This capacitor C103 is a commutating element to minimize the switching time of transistor Q25 and thereby reduce any overshoot of the ramp. For example, when the inverting amplifier 300 switches to ground potential, the transistor Q25 will then shut off causing the integrating amplifier 302 to ramp toward ground. Node 306 will remain at ground potential until the amplifier 300 switches to a high state when the negative ramp falls below the new threshold to cause the switching to a positive ramp. If reference will now be directed to FIG. 12, the detailed circuitry for generating the altitude compensation correction of the system will be more fully explained. Basically, altitude compensation circuit includes a two-bit binary counter that is reset to 00 and then counts 10, 01 and 11. The counter is formed by two J-K bistables 700, 702 connected such that a positive going trigger to the C input of the flip-flop 700 will cause an increment in the count. Thus, as is conventional, the J input of the bistable 700 is tied to a power line 703 and likewise the J and K inputs of the bistable 702. The power line 703 is connected to the positive supply +A through a diode CR700. The C input of bistable 702 is connected to the Q output of bistable 700 and the K input of flip-flop 700 is connected to the Q output of the bistable 702. The direct resets, R, of both bistables are connected to a reset line 706 which will reset the counter to 00 on a positive going transition on the reset line. Before the automobile is started, the key in the ignition is switched to the on position and power coming on in the circuit will produce a positive level at the output of an amplifier 708. The amplifier is initially on in a high condition because the noninverting input receives full supply voltage +A via the serial combination of a capacitor C702 and a resistor R721. The output of the amplifier 708 will remain high until the capacitor C702 charges thereby lowering the voltage to the noninverting input to the amplifier 708 below a threshold set at the inverting input. The threshold for the inverting input is supplied by the junction voltage of a divider resistor R716 and a divider resistor R717 being connected between the source of positive voltage +A and ground. The positive going edge of this initial pulse is used to reset the counter by its transmission to the reset line 706 via the junction of a capacitor C703 and a resistor R724 whose other terminals are connected between the output of the amplifier 708 and ground, respectively. The positive going edge of the pulse output from the amplifier 708 becomes a positive spike after being differentiated by the capacitor and resistor combination to reset the counter so that the Q outputs of the counter will initially read 00. This will begin a sampling process to determine the altitude at which the engine is presently being operated. Further, the reset pulse from the output of amplifier 708 is used to turn on a transistor Q700 via a resistor R708. The collector of transistor Q700 is connected to the junction of a capacitor C700 and a resistor R701 which are connected between a voltage node 720 an |